1. Field of the Invention
The present invention relates to apparatus and methods for communication of signals in a communication medium. More particularly, this invention relates to apparatus and methods to cancel echo interference and near-end crosstalk interference in a received signal from a transmitter attached to the communication system and from transmitters attached to other communication media in close proximity to the communication media.
2. Description of the Related Art
The gigabit Ethernet (1000 BASE-T) as defined by the IEEE standard 802.3ab is well known in the art. The structure capabilities and design consideration are described in:                “Gigabit Ethernet Over 4-Pair 100 OHM Category 5 Cabling,” Gigabit Ethernet Alliance, Cupertino, Calif., 1999,        “Gigabit Ethernet 1000 Base-T,” 1000 BASE-T Tutorial Series, Interoperability Laboratory Gigabit Ethernet Consortium, University of New Hampshire, Durham, N.H., 1998,        “Design Considerations for Gigabit Ethernet 1000 Base-T Twisted-pair Transceivers,” Hatamian et al., Proceedings of the IEEE 1999 Custom Integrated Circuit Conference, IEEE, 1998, pp. 335–342.        
Transmitting a gigabit data stream over four pair of category 5 unshielded twisted-pair cabling as described in the above-referenced papers has several design challenges. These challenges include signal attenuation, echo return loss, crosstalk characteristics of the cable, and electromagnetic emission and susceptibility.
Attenuation is the signal loss of the cabling from the transmitter to the receiver. Attenuation increases with frequency, which is due to such factors as skin effect. To minimize the effect of attenuation, the lowest possible frequency range that is consistent with the required data rate must be employed.
Echo is a by-product of the dual-duplex operation, where both the transmit and receive signal occupy the same wire pair. The residual transmit signal due to the trans-hybrid loss and the cabling return loss combine to produce an unwanted signal referred to here as echo.
Return loss is a measure of the amount of power reflected due to cabling impedance mismatches.
Crosstalk is an unwanted signal coupled between wire pairs that are in close proximity. Since 1000 BASE-T will use all four wire pairs, each pair is affected by crosstalk form the adjacent three pairs. Crosstalk is characterized in reference to the transmitter. Near-end crosstalk (NEXT) is crosstalk that appears at the output of a wire pair at the transmitter end of the cable and far-end crosstalk (FEXT) that appears at the output of a wire pair at the far end of the cable from the transmitter. Equal level far-end crosstalk (ELFEXT) is FEXT with the cable attenuation removed to provide equal level comparisons, i.e. crosstalk and receive signals voltages are compared at the end of the cabling opposite the transmitter. Crosstalk must be minimized to insure correct symbol recovery operations in the receiver.
A transmission system operating over unshielded cable must be capable of withstanding radiated energy from other sources, including AM, CB, short wave radio, and other external transmitters. The transmission system is required to have a tolerance to a 3 V/m continuous wave source above 27 MHz.
A further requirement is that the transmission system be immune to background and impulse noise. Impulse noise can be generated by power line transients, electrical fast transients, electrostatic discharge (ESD), and other sources.
Refer now to FIG. 1 for a discussion of near-end echo interference and near-end crosstalk. FIG. 1 shows a diagram of a gigabit Ethernet communications system. The gigabit Ethernet has two nodes that transmit and receive 1000 M bits per second (bps) full-duplex and bi-directionally. Each node consists of four transmitter/receivers (transceivers) 5a, 5b, 5c, 5d, 15a, 15b, 15c, and 15d that transmit 250 Mbps each.
Each transceiver 5a, . . . , 5d, 15a, . . . , 15d is connected to one end of one of four pair of unshielded twisted-pair cable 10a, 10b, 10c, and 10d. The transmitter 2 of each transceiver 5a, . . . , 5d, 15a, . . . , 15d forms a five level pulse amplitude modulated (PAM-5) shaped pulse signal that is transferred through the hybrid network 6 to one of the unshielded twisted-pair cable 10a, . . . , 10d. The transmitted signal traverses the unshielded twisted-pair cable 10a, . . . , 10d and is transferred through the hybrid network 6 to the receiver 4. The received signal is sensed, retimed, equalized and transferred to other circuitry for extraction of the digital data.
The full-duplex bi-directional transmission consists of transmitting and receiving data simultaneously in both directions on each of the four wire pairs, minimizing the symbol rate (and thus, the occupied signal bandwidth) on each wire pair by one half, as compared to unidirectional transmission and reception. The hybrid network 6 is used to enable bi-directional transmission over single wire pairs by filtering out the transmit signal at the receiver. The hybrid network 6 has good trans-hybrid loss to minimize the amount of transmitter signal that is coupled into the receiver 4, but it still cannot remove all of the transmitted signal from the adjacent transmitter 2. The residual transmitted signal from the adjacent transmitter 2 from the hybrid 6 is defined as the transmit echo signal 25.
Since the unshielded twisted-pair cable 10a, 10b, 10c, 10d are placed in close proximity to each other, often within the same cable, the crosstalk 20a, 20b, and 20c from transmitters within the same gigabit Ethernet node are coupled to the receiver 2. The near-end crosstalk 20a, 20b, and 20c and the transmit echo signal 25 must be cancelled from the receive signal to permit recovery of the transmitted signal.
Refer now to FIG. 2 for a more detailed discussion of the structure of a gigabit Ethernet node at one end of a bundled cable 10 containing four unshielded twisted-pair cable 10a, . . . , 10d. Each transmitter 4a, 4b, 4c, and 4d receives an encoded and scrambled symbol to be transmitted from the side-stream scrambler and symbol encoder 30, which in turn has received the digital data to be transferred from the Gigabit Media Independent Interface (GMII) 75. The digital transmit filter 35 shapes the symbol to be transmitted to condition the transmitted spectrum. The digital-to-analog converter 40 creates the five level pulse amplitude modulation signal that is then transferred to the hybrid network 6a for transmission on the unshielded twisted-pair cable 10a. 
A similar pulse amplitude modulated signal is simultaneously transmitted from a transmitter connected at the opposite end of the unshielded twisted-pair cable 10a. The received signal is separated from the transmitted signal in the hybrid network 6a and is the input to the analog-to-digital converter 45. The digitized received signal is the feed-forward equalizer (FFE) 50 to compensate for signal distortion introduced in the communication channel. The feed-forward equalizer 50 combined with a feedback equalizer or decision feedback equalizer 80 often provides better signal equalization than linear equalization when the transmission medium (cable 10) introduces strong signal attenuation with specific frequency regions. The feed-forward equalizer 50 does not modify the noise (echo, crosstalk, etc.).
The input signals from the side-stream scrambler and symbol encoder 30 to each transmitter 2b, 2c, and 2d are the inputs to the near-end crosstalk cancellers 55a, 55b, and 55c and to the echo canceller 60. The near-end crosstalk cancellers 55a, 55b, and 55c and the echo canceller 60 reproduce the near-end echo interference and the near-end crosstalk interference that is present in the received signal. The outputs of the near-end crosstalk cancellers 55a, 55b, and 55c and the echo canceller 60 are the inputs to the summing circuit 65. The equalized digitized received signal is transferred from the feed-forward equalizer 50 to the summing circuit 65. The summing circuit 65 combines the reproduction of the echo interference and the near-end crosstalk interference and the equalized digitized received signal to cancel the echo interference signal and the near-end crosstalk signals induced to the received signal as described above.
The Viterbi decoder and side-stream descrambler 70 provide error correction and resequencing of the receive signal to recover the digital data that is transferred to the Gigabit Media Independent Interface (GMII) 75 for further processing.
The near-end echo canceller 60 and the crosstalk cancellers 55a, 55b, and 55c are known in the art and have been applied to applications such as 100 Base-T Ethernet, asynchronous transfer mode (ATM), local area networks (LAN), and telephone communication networks.
U.S. Pat. No. 4,995,104 (Gitlin) describes a receiver that includes an interference canceller, which receives a corrupted signal and makes an estimate of the desired signal, subtracts the estimated desired signal from a delayed version of the received signal to form an estimate of the interference signal, then forms a final estimate of the desired signal by subtracting the estimated interference from a second delayed version of the received signal.
U.S. Pat. No. 5,329,586 (Agazzi) teaches an echo canceling circuit and associated method for canceling errors encountered in data communications decomposing a lookup-table nonlinear echo canceller into a plurality of smaller lookup tables, and combining outputs of the lookup tables.
U.S. Pat. No. 5,887,032 (Cioffi) discusses a method and apparatus for crosstalk cancellation (e.g., NEXT interference) from received signals on a line by adaptively estimating the crosstalk interference from the other lines having interfering transmissions and by canceling the crosstalk interference using the estimated crosstalk interference.
U.S. Pat. No. 4,669,116 (Agazzi) discloses an echo cancellation circuit for use with full-duplex data transmission systems. The echo canceller can operate in spite of time invariant non-linearities in the echo channel or in the implementation of the echo canceller itself (such as in D/A converters).
“A Pipelined Adaptive NEXT Canceller,” Im, et al., IEEE Transactions on Signal Processing, pp. 2252–2258, August 1998 Vol. 46 Issue: 8 ISSN: 1053-587X describes a near-end crosstalk (NEXT) canceller using a fine-grain pipelined architecture.
“100BASE-T2: 100 Mbit/S Ethernet Over Two Pairs Of Category-3 Cabling” Cherubini, et al., 1997 IEEE International Conference on Communications, pp. 1014–1018, 1997 Vol. 2, discusses the 100BASE-T2 physical layer specification for the receivers, particularly the adaptive digital filters that are required for echo and NEXT cancellation, equalization, and interference suppression.
As described above, near-end echo comes from the imperfections of the near-end hybrid network 6a that separates received signal from the transmitted signal and takes the significant part of the overall signal energy. The near-end echo signal degrades the receiver performance to a great extent and is a large source of error in the received signal. Therefore, the near-end echo canceller 60 is used to reduce the echo signal in the received signal. An adaptive echo cancellation technique is normally used because of its superior performance. For the receivers like gigabit Ethernet receivers, the performance requirement for the echo canceller is severe, and robustness of the received signal processing is required. Near-end crosstalk (NEXT), as described above, comes from the cross-coupling of the unshielded twisted-pair cable 10a, 10b, 10c, and 10d within a cable bundle and is one of the major sources of noise for the received signal.
In order to insure that the near-end echo and near-end crosstalk are minimized, a separate near-end echo/near-end crosstalk cancellation technique is necessary. The cancellation technique must not interact with other circuits in the receiver. In a receiver adopting a randomizing scrambler in the side-stream scrambler and symbol encoder 30, a correlator maybe used for the echo cancellation as shown in FIG. 3. A correlator 100 obtains the echo signal response for the transmitted symbols X(k) 115. Once the echo/NEXT signal response is acquired, a Finite Impulse Response (FIR) filter 105 set with the coefficients C0, . . . , Cj 125 obtained by the correlator 100 generates the duplicated echo e (k) 135 and subtracts the duplicated echo/NEXT signal e(k) 135 from the received signal X(k) 115 at the receiver. The coefficients C0, . . . , Cj 125 at the FIR 105 may be updated once they are set. However, the coefficients C0, . . . , Cj 125 at the FIR 105 are not usually updated once set because it usually takes thousands of symbols to generate new coefficients C0, . . . , Cj 125 from the received signal X(k) 115 in the correlator.
In a time varying echo/NEXT channel, such as the gigabit Ethernet, fixed coefficients C0, . . . , Cj 125 cannot serve the purpose of the echo/NEXT cancellation since they cannot reflect the changes of the channel characteristics. Therefore, an update method for the coefficients C0, . . . , Cj 125 is necessary. A windowed measurement for the coefficients C0, . . . , Cj 125 is possible, but a more preferred way is to update the coefficients C0, . . . , Cj 125 to reflect the most recent parameters of the channel.
Refer now to FIG. 4 for a discussion of the correlator 100. Each coefficient C0, . . . , Cj 125 of the FIR filter 105 is the normalized product of the received signal and a delayed version of the transmitted symbol b(k) 120. That isCJ(k)=N(x(k)*b(k−j))                where:                    Cj is the value of each coefficient 125 of the FIR filter 105.            N is a normalizing factor for the coefficients 125.            X(k) is the magnitude of the received signal 115.            b(k−j) is the magnitude of the delayed transmitted signal 120.                        
The multiplier circuit 150a, 150b, 150c, . . . , 150d receives the received signal X(k) 115 and the delayed transmitted signals b(k−j) 165a, 165b, 165c, . . . , 165d. The delayed transmitted signal b(k−j) 165a, 165b, 165c, . . . , 165d is the transmitted signal b(k) 120 successively delayed through each of the unit delay elements 150a, 150b, 150c, . . . , 150d. The product of the received signal X(k) 115 and the delayed transmitted signal 165a, 165b, 165c, . . . , 165d is the output of the multiplier 150a, 150b, 150c, . . . , 150d and the input of the normalization circuit 160a, 160b, 160c, . . . , 160d. The product is normalized to form the coefficients C0, C1, C2, . . . , Cj 125, by summing a large number of products and dividing the sum of the squares of the delayed transmitted signal b(k−j). Thus, each of the coefficients is determined as:       C    J    =                    ∑                  k          =          0                n            ⁢              (                              x            ⁡                          (              k              )                                *                      b            ⁡                          (                              k                -                j                            )                                      )                            ∑                  k          =          0                n            ⁢                        (                      b            ⁡                          (                              k                -                j                            )                                )                2                            where;                    n is the number of symbols used to determine the coefficient Cj.                        
In the conventional prior art, the number of symbols n used to determine the coefficient Cj is 1024. Therefore, for example, the first coefficient C0 becomes       C    0    =                    ∑                  k          =          0                1023            ⁢              (                              x            ⁡                          (              k              )                                *                      b            ⁡                          (              k              )                                      )                            ∑                  k          =          0                1023            ⁢                        (                      b            ⁡                          (              k              )                                )                2            